Controller of power converter

ABSTRACT

A controller of a power converter including an inverter that includes plural semiconductor switching elements. The controller suppresses an error between a voltage command and an inverter output voltage and responds to a voltage command at a high speed. The controller includes a voltage command generator that generates a voltage command signal and a switching pattern calculator that calculates and outputs, based on the voltage command signal, a switching pattern of a synchronous PWM system in which an average value of an inverter output voltage matches the voltage command signal.

TECHNICAL FIELD

The present invention relates to a controller of a power converter thathas a plurality of semiconductor switching elements, and moreparticularly to a synchronous pulse width modulation (hereinafter,referred to as “PWM”) control that synchronizes a voltage command, whichis output to a PWM inverter controlled by using a PWM, with a switchingpattern.

BACKGROUND ART

In a synchronous PWM control, a switching pattern to control a PWMinverter is calculated. As representative systems that calculate aswitching pattern, a system that synchronizes a carrier wave such as atriangular wave with a phase angle of a voltage command (hereinafter,“carrier-wave comparison system”) and a system directly referring to aphase of a voltage command (hereinafter, “phase reference system”) canbe exemplified. In these systems, the carrier-wave comparison system cansimplify a configuration of a control system and is excellent in theresponsiveness to a voltage command, whereas the phase reference systemcan effectively suppress a higher harmonic component contained in aninverter output voltage. Conventionally, there are many technicalliteratures about the carrier-wave comparison system. NonpatentLiteratures 1 and 2 and Patent Document 1 mentioned below arerepresentative literatures about the phase reference system.

When a synchronous PWM control is performed, an approximate shape of aswitching pattern can be recognized in many cases. This means that, inthe synchronous PWM control, the shape of an inverter output voltage canbe recognized in advance. Therefore, in the synchronous PWM control, aswitching phase can be obtained in advance to be able to obtain desiredcharacteristics of an inverter output-voltage waveform in one cycle of avoltage command.

Nonpatent Literatures 1 and 2 disclose setting methods of a switchingphase that make it possible to perform suppression of a higher harmoniccomponent contained in an inverter output voltage and assigning anarbitrary fundamental wave component. Patent Document 1 discloses asetting method of a switching phase in which a fundamental wavecomponent contained in an inverter output-voltage waveform matches avoltage command.

Patent Document 1: Japanese Patent Application Laid-open No. H6-253546

Nonpatent Literature 1: IEEE Transactions On Industry Applications(1973, Vol.IA-9, No. 3), Generalized Techniques of Harmonic Eliminationand Voltage Control in Thyristor Inverters Part I-Harmonic Elimination

Nonpatent Literature 2: IEEE Transactions On Industry Applications(1974, Vol. 10, No. 5), Generalized Techniques of Harmonic Eliminationand Voltage Control in Thyristor Inverters Part II-Voltage ControlTechniques

DISCLOSURE OF INVENTION Problem to be Solved by the Invention

While characteristics of the carrier-wave comparison system and thephase reference system as representative synchronous PWM control systemshave been explained above, these control systems have the followingtechnical problems.

In the carrier-wave comparison system, as far as an amplitude and aphase of a fundamental wave component of an inverter output voltage areconcerned, there is a problem such that a relatively large error occursbetween the amplitude and a voltage command although the phase matches aphase of the voltage command. This error is thought to have thefollowing influences.

(1) When a motor as a load is controlled by applying an open loopcontrol system such as a V/f control, for example, motor torqueprecision decreases due to excess or lack of an inverter voltage output.(2) When a current control of a motor as a load is performed, forexample, a current control gain varies equivalently.(3) When a control of substituting an inverter output voltage with avoltage command is performed, a voltage limiter process or the like isinfluenced, and thus a current control system becomes unstable.

Therefore, in the carrier-wave comparison system, gain compensation orthe like is performed for a voltage command.

Meanwhile, in the phase reference systems described in NonpatentLiteratures 1 and 2 and Patent Document 1 mentioned above, theresponsiveness to a voltage command decreases. For example, when acurrent control of a motor as a load is performed, a voltage commandvaries minutely in order to flow a predetermined current. In the phasereference systems described in Nonpatent Literatures 1 and 2 and PatentDocument 1, a switching phase of a switching pattern to obtain desiredcharacteristics is calculated by using a Fourier analysis or the like.Therefore, it is a common procedure that a switching phase of aswitching pattern in a control system is expressed by a function or atable of a voltage command amplitude. Besides, in accordance with thevariation of the voltage command, the switching phase also variesminutely, and a switching phase set to obtain desired characteristics isnot reproduced. Consequently, a priority control of a switching phasebecomes necessary. However, when a control of prioritizing a switchingphase set in advance is performed, a reflection of a voltage-commandamplitude-change to a switching phase is limited to one time per onecycle or per a half cycle of a voltage command. This lowers theresponsiveness to the voltage command.

In summary, the carrier-wave comparison system has a problem such that arelatively large error occurs between a voltage command and afundamental wave component of an inverter output voltage, although theresponsiveness to a change of a voltage command becomes relatively fast.The phase reference system has a problem such that the responsiveness toa voltage command is lowered, particularly when the system is to obtaindesired characteristics by a switching phase set by using Fourieranalysis.

The present invention has been made in view of the above, and an objectof the invention is to provide a controller of a power converter capableof suppressing an error between a voltage command and an inverter outputvoltage and capable of responding to a voltage command at a high speed,even when a phase reference system is applied.

Means for Solving Problem

To solve the problem described above and achieve the object, there isprovided a controller of a power converter according to the presentinvention, applied to a power converter having an inverter that has aplurality of switching elements and controls the switching elements ofthe inverter by using a pulse width modulation, wherein the controllerincludes: a voltage-command generator that generates a voltage commandsignal; and a switching pattern calculator that calculates a switchingpattern to control the semiconductor switching elements of the inverterbased on the voltage command signal, and the switching patterncalculator calculates a switching pattern of a synchronous PWM system,and outputs a switching pattern in which an average value (anoutput-voltage average value) of output voltages output from theinverter matches the voltage command.

EFFECT OF THE INVENTION

According to the controller of a power converter of the presentinvention, a switching pattern calculator calculates a switching patternof a synchronous PWM system, and outputs a switching pattern in which anaverage value of inverter output voltages matches a voltage command.Therefore, even when a phase reference system is applied, the controllercan suppress an error between a voltage command and an inverter outputvoltage and can respond to a switching command at a high speed.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 depicts a basic configuration of a power converter according to afirst embodiment of the present invention.

FIG. 2 is a block diagram of a functional configuration of a controllerof a power converter according to the first embodiment.

FIG. 3 depicts a relationship between a voltage command vector input toa switching pattern calculator and a dq coordinate system of each signalprocessed by the switching pattern calculator.

FIGS. 4( a) to 4(d) are explanatory diagrams of an operation of thecontroller according to the first embodiment.

FIG. 5 is a table for classifying a switching operation of an invertercontrolled by the controller according to the first embodiment by aphase timing.

FIGS. 6( a) to 6(d) are explanatory diagrams of an operation of acontroller according to a second embodiment.

FIGS. 7( a) to 7(d) depict enlarged sections A to G shown in FIG. 6.

FIG. 8 is a table for classifying a switching operation in a synchronousfive-pulse mode by a phase timing.

FIGS. 9( a) to 9(c) are explanatory diagrams of an operation of acontroller according to a fourth embodiment.

EXPLANATIONS OF LETTERS OR NUMERALS

-   -   10 Power converter    -   21 Direct-current power source    -   22 Inverter    -   221 Semiconductor switching element (U-phase P-side)    -   222 Semiconductor switching element (V-phase P-side)    -   223 Semiconductor switching element (W-phase P-side)    -   224 Semiconductor switching element (U-phase N-side)    -   225 Semiconductor switching element (V-phase N-side)    -   226 Semiconductor switching element (W-phase N-side)    -   23 Load    -   50 Control unit    -   51 Voltage command generator    -   52 Voltage command signal (d-axis on biaxial        orthogonal-rotational-coordinate)    -   53 Voltage command signal (q-axis on biaxial        orthogonal-rotational-coordinate)    -   54 Switching pattern calculator    -   541 Phase calculator    -   542 Phase signal (on dq coordinate system)    -   544 Voltage-command phase signal (U-phase)    -   546 Voltage-command norm signal    -   548 Sampled-held voltage-command norm signal    -   543 Adder    -   545 Norm calculator    -   547 Sampling and holding (S/H) unit    -   549 Switching phase calculator    -   55 Coordinate-conversion phase signal    -   550 Switching phase signal    -   551 Phase comparing unit    -   56 Switching pattern signal

BEST MODE(S) FOR CARRYING OUT THE INVENTION

Exemplary embodiments of a controller of a power converter according tothe present invention will be explained below in detail with referenceto the accompanying drawings. The present invention is not limited tothe embodiments.

First Embodiment

FIG. 1 depicts a basic configuration of a power converter according to afirst embodiment of the present invention. As shown in FIG. 1, a powerconverter 10 is configured to include a direct-current power source 21,an inverter 22, and a control unit 50 that controls semiconductorswitching elements 221 to 226 of the inverter 22 by using a PWM. Thepower converter 10 is connected to a load 23. The direct-current powersource 21 supplies a direct-current power to the inverter 22. Theinverter 22 includes the semiconductor switching elements 221 to 223 assemiconductor switching elements at a P side, and the semiconductorswitching elements 224 to 226 as semiconductor switching elements at anN side. The inverter 22 forms a series circuit having a seriesconnection between the semiconductor switching element 221 as asemiconductor switching element at the P side and the semiconductorswitching element 224 as a semiconductor switching element at the Nside. Both ends of this series circuit are connected to positive andnegative power source terminals of the direct-current power source 21. Arelationship between the semiconductor switching element 222 and thesemiconductor switching element 225, and a relationship between thesemiconductor switching element 223 and the semiconductor switchingelement 226 are the same, and both ends of each series circuit are alsoconnected to the positive and negative power source terminals of thedirect-current power source 21. FIG. 1 depicts a configuration of atwo-level three-phase inverter, as an example; however, theconfiguration of the inverter is not limited thereto, and a powerconverter including an inverter other than the two-level three-phaseinverter can be also used.

FIG. 2 is a block diagram of a functional configuration of thecontroller of a power converter according to the first embodiment, anddepicts details of a configuration of the control unit 50 shown inFIG. 1. The control unit 50 includes a voltage command generator 51 anda switching pattern calculator 54. The switching pattern calculator 54includes a phase calculator 541, an adder 543, a norm calculator 545, asampling and holding unit (hereinafter “S/H unit”) 547, a switchingphase calculator 549, and a phase comparing unit 551.

An operation of the controller according to the first embodiment isexplained next with reference to FIG. 2 and FIG. 3. FIG. 3 depicts arelationship between a voltage command vector input to the switchingpattern calculator 54 and a biaxial orthogonal-rotational-coordinatesystem (hereinafter, “dq coordinate system”) of each signal processed bythe switching pattern calculator 54.

In FIG. 2, the voltage command generator 51 outputs voltage commandsignals 52 and 53 in the dq coordinate system to the switching patterncalculator 54. The voltage command signal 52 is a voltage commandcomponent in a d-axis direction, and the voltage command signal 53 is avoltage command component in a q-axis direction. The input voltagecommand signals 52 and 53 are then input to the phase calculator 541 tocalculate a phase signal 542. The phase calculator 541 is a functionalunit that performs an arc tangent calculation. The phase signal 542calculated by the phase calculator 541 and the input voltage commandsignals 52 and 53 have a relationship shown in FIG. 3.

When the voltage command signal 52 is Vd*, the voltage command signal 53is Vq*, and the phase signal 542 is θv, a relationship of the followingequation is established. The phase calculator 541 can obtain the phasesignal 542 by directly calculating this equation or by referring to atable prepared in advance.

$\begin{matrix}{{\theta \; v} = {\arctan \left( \frac{{Vq}^{*}}{{Vd}^{*}} \right)}} & \left( {1\text{-}1} \right)\end{matrix}$

The adder 543 adds the phase signal 542 to a coordinate-conversion phasesignal 55, and obtains a voltage-command phase signal 544 on a two-phasestationary coordinate system (hereinafter, “αβ coordinate system”). Theadder 543 performs this adding process and a process of accommodatingadded phase signals within a range of 0π to 2π. The norm calculator 545calculates a voltage-command norm signal 546 from the voltage commandsignals 52 and 53. FIG. 3 also depicts a relationship between thevoltage-command norm signal 546 and other signals.

When the voltage-command norm signal 546 is Vn*, a relationship of thefollowing equation is established. The norm calculator 545 can obtainthe voltage-command norm signal 546 by directly calculating thisequation or by referring to a table, in a similar manner to that ofobtaining the phase signal 542.

Vn*=√{square root over ((Vd*)²+(Vq*)²)}{square root over((Vd*)²+(Vq*)²)}  (1-2)

The S/H unit 547 samples and holds the voltage-command norm signal 546obtained by the norm calculator 545, and inputs this signal to theswitching phase calculator 549. The switching phase calculator 549calculates a switching phase signal 550. The phase comparing unit 551outputs a switching pattern signal 56 by referring to thevoltage-command phase signal 544 and the switching phase signal 550. Theswitching pattern signal 56 is output to the inverter 22. That is, eachsemiconductor switching element is controlled in accordance with theswitching pattern signal 56.

The switching phase signal 550 and the switching pattern signal 56 areshown by plural arrows in FIG. 2. These signals correspond to controlsignals to semiconductor switching elements of the inverter 22. That is,the number of outputs of the switching phase signal 550 and theswitching pattern signal 56 changes depending on the number of phases ofthe power converter and a kind of the number of levels.

An operation of the switching phase calculator 549 is explained next.The switching phase calculator 549 calculates the switching phase signal550 from the voltage-command norm signal 546 in this example. An indexcalled an average value of output voltages output from the inverter 22(hereinafter, simply “output-voltage average value”) is introduced as anevaluation index of a switching pattern calculation. High precision ofan output voltage can be achieved by providing a switching pattern inwhich this output-voltage average value matches a voltage command.

Preferably, the output-voltage average value and the voltage command arevalues in the dq coordinate system. This is because the dq coordinatesystem as a rotational coordinate can take into account a phase changefollowing a time progress, in considering the output-voltage averagevalue. By this control, an error generated when comparing a phase changewith the average value in the αβ coordinate system can be suppressed,and a phase delay of an inverter output voltage can be suppressed as aresult of this control.

A switching pattern calculation can be simplified by using theoutput-voltage average value as a component of the dq coordinate systemin a voltage-command vector direction. Conversely, when theoutput-voltage average value is not a component in the voltage-commandvector direction, an average value in each of the d-axis component andthe q-axis component needs to be considered. However, both cannot besimultaneously satisfied sometimes in the switching pattern calculation,and in such a case, an operation of setting priorities of the bothcomponents is necessary. This kind of calculation can be omitted byusing a component in the voltage-command vector direction. In thesynchronous PWM control, a switching pattern is output synchronouslywith a voltage command phase of the αβ coordinate system. Therefore, theoutput-voltage average value is calculated preferably based on a phaseof the αβ coordinate system.

Operations of the switching phase calculator 549 and the phase comparingunit 551 are explained next with reference to FIG. 4 and FIG. 5. FIGS.4( a) to 4(d) are explanatory diagrams of an operation of the controlleraccording to the first embodiment, and FIG. 5 is a table for classifyinga switching operation of an inverter controlled by the controlleraccording to the first embodiment by a phase timing. For the sake ofexplanation, a two-level three-phase inverter is taken as an example,and a case that this inverter is controlled in a synchronous three-pulsemode is explained.

In FIG. 4, FIG. 4( a) depicts a time in a lateral axis, and depicts aphase of a U-phase voltage command (a U-phase voltage-command phase) ina vertical axis. FIGS. 4( b) to 4(d) depict a time in a lateral axis, aP-side switching pattern in each phase in a vertical axis, and inverteroutput voltages, respectively. As shown in FIG. 4( a), time and aU-phase voltage command are in a proportional relationship, andtherefore FIGS. 4( b) to 4(d) can be regarded as a relationship withrespect to the U-phase voltage-command phase.

FIGS. 4( c) and (d) depict waveforms of an output voltage of theinverter observed on the dq coordinate system. FIG. 4( c) depicts acomponent in a voltage-command vector direction (hereinafter,“voltage-command-vector direction component”) and FIG. 4( d) depicts awaveform of a component in a direction orthogonal to the voltage-commandvector direction (hereinafter, “voltage-command-vectororthogonal-direction component”). Although a voltage command waveform ofa U-phase is not shown in FIG. 4, it can be obtained by a cosinecalculation of a phase in FIG. 4( a).

As shown in FIG. 4, when a two-level three-phase inverter is controlledin a synchronous three-pulse mode, switching occurs 18 times in onecycle of a voltage command phase. In FIG. 4( b), switching can beclassified into a switching group in which a phase timing changesdepending on a size of a voltage command (hereinafter, “i group”) and aswitching group in which a phase timing does not change (hereinafter,“ii group”). For the sake of explanation, numbers (1) to (18) are givento switching operation points (hereinafter, simply “operation point”)(see FIG. 4( b) and FIG. 5). As far as operation points of the ii groupand an intermediate point of these operation points are concerned, asection can be divided into 12 sections including sections A to L asshown in FIG. 4( b).

In these sections, a start or an end becomes an operation point, andeach section necessarily includes operation points of the i group at oneposition. Therefore, these sections become a minimum section capable ofcontrolling the output-voltage average value, for the following reasons.In each section defined above, while operation points of the ii groupare fixed points to determine a start or an end of each section,operation points of the i group are changeable within each section.

As shown in FIG. 4( b), Δθ is introduced as a parameter to determine aswitching timing in the section A. When this Δθ is used, a phase timingof each switching takes a value as shown in FIG. 5. Each of these valuescorresponds to the switching phase signal 550 output from the switchingphase calculator 549 (see FIG. 2).

That is, in each section, a timing of switching in the section A (phase:Δθ) is controlled such that the voltage-command-vector directioncomponent of the inverter output shown in FIG. 4( c) matches the voltagecommand. For example, in a section starting at an operation point (2),that is, the section B, a timing control becomes possible at anoperation point (3) by manipulating Δθ.

In the section B, before the operation point (3), switching in phases U,V, and W becomes “on”, “on”, and “off”, respectively. When a load isbalanced, an output voltage in each phase of the inverter can beexpressed by the following equations, where Vdc denotes an outputvoltage of the direct-current power source 21.

$\begin{matrix}{{Vu} = {\frac{1}{3}{Vdc}}} & \left( {1\text{-}3} \right) \\{{Vv} = {\frac{1}{3}{Vdc}}} & \left( {1\text{-}4} \right) \\{{Vw} = {{- \frac{2}{3}}{Vdc}}} & \left( {1\text{-}5} \right)\end{matrix}$

When the value is converted into a value on the αβ coordinate system, itis expressed by the following equations.

$\begin{matrix}{{V\; \alpha} = {\frac{1}{2}\sqrt{\frac{2}{3}}{Vdc}}} & \left( {1\text{-}6} \right) \\{{V\; \beta} = {\frac{1}{\sqrt{2}}{Vdc}}} & \left( {1\text{-}7} \right)\end{matrix}$

These are then coordinate-converted into values on a rotationalcoordinate. When the U-phase voltage-command phase shown in FIG. 4( a)is used, a vector component can be divided into a voltage-command-vectordirection component (hereinafter, “dv-axis”) and avoltage-command-vector orthogonal-direction component (hereinafter,“qv-axis”). FIG. 3 depicts details of the dv-axis and the qv-axis.

When the U-phase voltage-command phase shown in FIG. 4( a) is θvu,voltages can be expressed by the following equations, where θvucorresponds to the voltage-command phase signal 544 output from theadder 543 (see FIG. 2).

$\begin{matrix}{{Vdv} = {\sqrt{\frac{2}{3}}{{Vdc} \cdot {\sin \left( {{\theta \; {vu}} + \frac{\pi}{6}} \right)}}}} & \left( {1\text{-}8} \right) \\{{Vqv} = {\sqrt{\frac{2}{3}}{{Vdc} \cdot {\cos \left( {{\theta \; {vu}} + \frac{\pi}{6}} \right)}}}} & \left( {1\text{-}9} \right)\end{matrix}$

To calculate an average voltage in the dv-axis, integration is performedin each section, and a result of the integration is divided by a phase.This dv-axis average value is controlled to match the voltage commandnorm Vn*.

For example, in the section B, control is performed to establish thefollowing equation. This equation takes into account a fact that avoltage is zero in a phase after the operation point (3).

$\begin{matrix}{{Vn}^{*} = {\frac{6}{\pi}{\int_{\pi/6}^{{\pi/3} - {\Delta\theta}}{\left\{ {\sqrt{\frac{2}{3}}{{Vdc} \cdot {\sin \left( {{\theta \; {vu}} + \frac{\pi}{6}} \right)}}} \right\} {\theta}}}}} & \left( {1\text{-}10} \right)\end{matrix}$

When the equation (1-10) is solved, Δθ is expressed by the followingequation. This Δθ can be calculated each time, or can be prepared in atable that corresponds to the voltage command norm Vn*.

$\begin{matrix}{{\Delta\theta} = {\sin^{- 1}\left( {\frac{1}{2} - {\sqrt{\frac{3}{2}} \cdot \frac{{Vn}^{*}}{Vdc} \cdot \frac{\pi}{6}}} \right)}} & \left( {1\text{-}11} \right)\end{matrix}$

While the section B is explained in the equation (1-11), the aboveexplanation is similarly applied to other sections as well. Waveforms inother sections are either the same as the waveform in the section B orare bilaterally symmetrical. Therefore, AO can be calculated by usingthe equation (1-11) or by an equation equivalent thereto.

Although the order of explanation is opposite, the switching phasecalculator 549 obtains AO following the equation (1-11) from avoltage-command norm signal 548 (Vn*), and outputs the switching phasesignal 550 as shown in FIG. 5. The phase comparing unit 551 refers tothe switching phase signal 550 and the voltage-command phase signal 544,and calculates the switching pattern signal 56 to be given to eachphase, as shown in FIGS. 4( a), 4(b), and FIG. 5.

In the configuration of the controller shown in FIG. 2, the S/H unit 547is not necessarily an essential component. For example, when a voltagecommand varies minutely such as when the voltage command generator 51performs a current control, a phenomenon called chattering occurs andthere is a risk that a switching operation is repeated for many times.The S/H unit 547 is effective to prevent such chattering and cancontribute to a stable operation of the power converter.

When a sampling and holding timing of the S/H unit 547 is at a boundaryof a section for calculating the output-voltage average value shown inFIG. 4, for example, this is convenient because the timing is consistentwith updating of the inverter output voltage. A sampling and holdingtiming other than the above can be also suitably set by matching a loadand a control mode of the voltage command generator 51. For example,when sampling and holding finer than the above timing is performed,waste of time can be suppressed and the responsiveness can be improved.

As explained above, according to the controller of a power converter ofthe present embodiment, the controller calculates and outputs aswitching pattern in which an output-voltage average value matches avoltage command. Therefore, an error between the voltage command and theinverter output voltage can be suppressed, and a voltage in highprecision can be obtained, even when a synchronous PWM control isapplied.

According to the controller of a power converter of the presentembodiment, the output-voltage average value using an evaluation indexof calculating a switching pattern is calculated by using a value on thedq-coordinate system. Therefore, a phase delay of an output voltage canbe suppressed.

According to the controller of a power converter of the presentembodiment, a voltage-command-vector direction component is used for anoutput-voltage average value. Therefore, calculation of a switchingpattern can be simplified.

Further, according to the controller of a power converter of the presentembodiment, the controller uses an average value of output-voltage insections obtained by dividing the voltage command phase into pluralsections, for an output-voltage average value. Therefore, a response tothe voltage command can be performed at a high speed.

As explained above, according to the controller of a power converter ofthe first embodiment, it is possible to effectively achieve thecompatibility of the precision of voltage commands and responsiveness,which conventional synchronous PWM control systems do not have.

Second Embodiment

In the first embodiment, a case of controlling a two-level three-phaseinverter in a synchronous three-pulse mode has been described as anexample. When control is performed in other pulse modes, calculation ofa switching pattern can be also performed in a similar manner to that ofthe first embodiment.

FIG. 6( a) depicts a U-phase voltage-command phase in a similar mannerto that in FIG. 4( a). Meanwhile, FIGS. 6( b) to 6(d) depict a P-sideswitching pattern in each phase when the two-level three-phase inverteris controlled in a synchronous five-pulse mode, and inverter outputvoltages in this case. As shown in these drawings, in the synchronousfive-pulse mode, a switching operation is performed 30 times in onecycle of a voltage command, and the voltage command phase is dividedinto 24 parts. For the sake of explanation, numbers (1) to (30) aregiven to operation points, and symbols from A to X are given torespective sections.

An operation of a controller according to a second embodiment isexplained next with reference to FIG. 7 and FIG. 8. FIGS. 7( a) to 7(d)depict sections A to G shown in FIG. 6 in an enlarged manner, and FIG. 8is a table for classifying a switching operation in the synchronousfive-pulse mode by a phase timing. Explanations of operations are givenby focusing on the section C and the section D.

In FIG. 7, inverter output-voltage waveforms in thevoltage-command-vector direction components in the section C and thesection D are different from those in the synchronous three-pulse modeof the first embodiment (see FIG. 4). As is clear from a comparisonbetween FIG. 7( b) and FIG. 4( b), two kinds of Δθ, that is, Δθ1 andΔθ2, are necessary to determine a timing in the synchronous five-pulsemode of the second embodiment. When these Δθ1 and Δθ2 are used, a phasetiming of each switching takes a value as shown in FIG. 8, and thesevalues correspond to the switching phase signal 550 output from theswitching phase calculator 549 (see FIG. 2).

In the section C and the section D, P-side switching states in the U, V,and W phases are “on”, “on”, and “on”, respectively, after an operationpoint (4) and before an operation point (5). In this case, becauseN-side switching states in the U, V, and W phases are “off”, “off”, and“off”, these sections become zero voltage sections. In contrast, P-sideswitching states in the U, V, and W phases are “on”, “on”, and “off”,respectively in other than these zero voltage sections, and are the sameas that in the section B explained in the first embodiment.

Therefore, inverter output-voltage waveforms in other than the zerovoltage sections can be expressed by the equation (1-8) and the equation(1-9) in the dv-axis direction and the qv-axis direction, respectively.Accordingly, in the section C, the following equation taking intoaccount the zero voltage section is calculated so that an average valuein the dv-axis direction matches the voltage command norm Vn*.

$\begin{matrix}{{Vn}^{*} = {\frac{12}{\pi}{\int_{\pi/6}^{{\pi/4} - {{\Delta\theta}\; 1}}{\left\{ {\sqrt{\frac{2}{3}}{{Vdc} \cdot {\sin \left( {{\theta \; {vu}} + \frac{\pi}{6}} \right)}}} \right\} {\theta}}}}} & \left( {2\text{-}1} \right)\end{matrix}$

Similarly, in the section D, the following equation taking into accountthe zero voltage section is calculated so that an average value in thedv-axis direction matches the voltage command norm Vn*.

$\begin{matrix}{{Vn}^{*} = {\frac{12}{\pi}{\int_{{\pi/4} + {{\Delta\theta}\; 2}}^{\pi/3}{\left\{ {\sqrt{\frac{2}{3}}{{Vdc} \cdot {\sin \left( {{\theta \; {vu}} + \frac{\pi}{6}} \right)}}} \right\} {\theta}}}}} & \left( {2\text{-}2} \right)\end{matrix}$

When the equation (2-1) and the equation (2-2) are solved, Δθ1 and Δθ2are expressed in the following equations. These Δθ1 and Δθ2 can becalculated each time, or can be prepared in a table that corresponds tothe voltage command norm Vn*, in a similar manner to that of the firstembodiment.

$\begin{matrix}{{{\Delta\theta}\; 1} = {{\frac{5}{12}\pi} - {\cos^{- 1}\left( {\frac{1}{2} - {\frac{\pi}{12}{\sqrt{\frac{3}{2}} \cdot \frac{{Vn}^{*}}{Vdc}}}} \right)}}} & \left( {2\text{-}3} \right) \\{{{\Delta\theta}\; 2} = {{{- \frac{5}{12}}\pi} + {\cos^{- 1}\left( {\frac{\pi}{12}{\sqrt{\frac{3}{2}} \cdot \frac{{Vn}^{*}}{Vdc}}} \right)}}} & \left( {2\text{-}4} \right)\end{matrix}$

While the section C and the section D have been explained with theequation (2-3) and the equation (2-4), the above explanation issimilarly applied to other sections. Specifically, in the switchingphase shown in FIG. 8, an output-voltage average value can match avoltage command by performing a switching control shown in FIG. 6( b).

As for the section for calculating an output-voltage average value, inthe two-level three-phase inverter explained in the first embodiment,this section becomes a voltage command phase divided by “6n−6”, where nis a number of synchronous pulses. That is, the number of times ofswitching a semiconductor switching element increases based on anincrease of the number of synchronous pulses, and an operable amount(degree of freedom) is expressed in addition to the amplitude and phaseof an output voltage. In Nonpatent Literatures 1 and 2 mentioned above,the degree of freedom is used to reduce higher harmonics. In the presentembodiment, the degree of freedom is used to increase the number oftimes of updating an inverter output voltage. In this respect,utilization of the degree of freedom is very different from that inNonpatent Literatures 1 and 2.

Third Embodiment

In the first embodiment, an embodiment in which, a voltage-command phasesection for calculating an output-voltage average value is divided into12 parts when a two-level three-phase inverter is controlled in asynchronous three-pulse mode, has been described as an example. In thesecond embodiment, an embodiment in which a voltage-command phasesection for calculating an output-voltage average value is divided into24 parts when the two-level three-phase inverter is controlled in thesynchronous five-pulse mode, has been described as an example. Incontrast, a third embodiment is an embodiment in which these numbers ofdivision are set to a half. That is, by setting two adjacent sections asa new section, the number of sections is reduced and the calculationtime and processing time are shortened.

As a concept of setting two adjacent sections as a new section, itsuffices that the following two conditions are satisfied:

(1) an inverter output voltage in a qv-axis is zero; and(2) when a point at which an output voltage becomes zero is a boundarypoint of sections, waveforms of adjacent sections are symmetrical abouta point. For example, in the embodiment shown in FIG. 4, the above twoconditions are satisfied at a boundary point between the section B andthe section C, as shown in FIG. 4( d). Therefore, the section A and thesection B are set as one section, and the section C and the section Dare also set as one section. In this way, “A, B”, “C, D”, “E, F”, “G,H”, “I, J”, and “K, L” become new sections in voltage phase sections inone cycle of a voltage command phase. The same Δθ can be used in thesesections.

When the above control is performed, the number of times of updating anoutput-voltage average value is reduced, and thus the responseperformance decreases as a result. However, an average value of avoltage-command vector orthogonal component (a qv-axis component) in theoutput-voltage average value can be set to zero, and therefore theprecision of an output voltage can be improved. This point can beexplained as follows.

Controlling a power converter by a two-level three-phase inverter in thesynchronous three-pulse mode is taken as an example in a similar mannerto that explained above. For example, in FIG. 4( b), in a section ABhaving combined the section A and the section B, a voltage in theqv-axis is calculated. The calculation order is the same as thatdescribed in the first embodiment, and therefore detailed explanationsthereof will be omitted. A voltage can be expressed by the followingequation (3-1), in a section before the operation point (2) (an originalsection A). However, this equation gives a value in a phase after theoperation point (1), and a qv-axis voltage becomes zero in a phasebefore the operation point (1). Further, a voltage can be expressed bythe following equation (3-2) in a section after the operation point (2)(an original section B). In this case, the qv-axis voltage becomes zeroin a phase after the operation point (3).

$\begin{matrix}{{Vqv} = {{- \sqrt{\frac{3}{2}}}{{Vdc} \cdot {\sin \left( {\theta \; {vu}} \right)}}}} & \left( {3\text{-}1} \right) \\{{Vqv} = {{- \sqrt{\frac{3}{2}}}{{Vdc} \cdot {\cos \left( {{\theta \; {vu}} + \frac{\pi}{6}} \right)}}}} & \left( {3\text{-}2} \right)\end{matrix}$

Next, an average value is calculated from the equation (3-1) and theequation (3-2). When a zero voltage section is considered, anoutput-voltage average value in the qv-axis direction (Vqv_AV) isderived by an equation (3-3) in the section A, and is derived by anequation (3-4) in the section B.

$\begin{matrix}\begin{matrix}{{Vqv\_ AV} = {\frac{6}{\pi}{\int_{\Delta\theta}^{\pi/6}{\left\{ {{- \sqrt{\frac{2}{3}}}{{Vdc} \cdot {\sin \left( {\theta \; {vu}} \right)}}} \right\} {\theta}}}}} \\{= {\frac{6}{\pi}{\sqrt{\frac{2}{3}} \cdot {Vdc} \cdot \left( {\frac{\sqrt{3}}{2} - {\cos ({\Delta\theta})}} \right)}}}\end{matrix} & \left( {3\text{-}3} \right) \\\begin{matrix}{{Vqv\_ AV} = {\frac{6}{\pi}{\int_{\pi/6}^{{\pi/3} - {\Delta\theta}}{\left\{ {{- \sqrt{\frac{3}{2}}}{{Vdc} \cdot {\cos \left( {{\theta \; {vu}} + \frac{\pi}{6}} \right)}}} \right\} {\theta}}}}} \\{= {\frac{6}{\pi}{\sqrt{\frac{2}{3}} \cdot {Vdc} \cdot \left( {{\cos ({\Delta\theta})} - \frac{\sqrt{3}}{2}} \right)}}}\end{matrix} & \left( {3\text{-}4} \right)\end{matrix}$

As explained above, average values of inverter output voltages in theqv-axis direction in the section A and the section B are the same exceptfor their polarities. Therefore, when Δθ in both equations are the same,the average value of inverter output voltages in the section AB becomeszero.

As explained above, according to the controller of a power converter ofthe present embodiment, the average value of the voltage-command vectororthogonal component (the qv-axis component) in the output-voltageaverage value can be set to zero. Therefore, the precision of an outputvoltage of the power converter can be improved.

Fourth Embodiment

In the synchronous PWM control explained in the first to thirdembodiments, examples of embodiments in the same pulse mode such as asynchronous three-pulse mode or a synchronous five-pulse mode have beendescribed. In the present embodiment, an embodiment in different pulsemodes, that is, a combination of pulse modes in different numbers ofsynchronous pulses, is described. Specifically, this embodiment is basedon a concept that a switching state in each phase does not change beforeand after performing switching between synchronous pulse modes, and thatthere is no negative influence when switching is freely performed at aboundary point between sections for calculating a voltage average valueexplained in the first to third embodiments.

FIGS. 9( a) to (c) are explanatory diagrams of an operation of acontroller according to the fourth embodiment. FIG. 9( b) depictsswitching patterns in phases in the synchronous three-pulse mode shownin FIG. 4, and FIG. 9( c) depicts switching patterns in phases in thesynchronous five-pulse mode shown in FIG. 6. In these drawings, indexes“3” and “5” are attached to distinguish between a section in thesynchronous three-pulse mode and a section in the synchronous five-pulsemode.

In FIG. 9, a boundary point between a section A3 and a section B3 is aboundary point (an operation point in the ii group) in sections forcalculating a voltage average value explained in the first to thirdembodiments, and a switching state in each phase does not change betweenpulse modes before and after this boundary point. Therefore, thisboundary point can be used as a switching timing of both pulse modes.Similarly, each boundary point between “a section C3 and a section D3”,“a section E3 and a section F3”, “a section G3 and a section H3”, “asection 13 and a section J3”, and “a section K3 and a section L3” can bealso used as a switching timing. That is, there are a plurality ofswitchable timings in one cycle of a voltage command. Therefore, when acontrol is performed by using a power converter as a two-levelthree-phase inverter and by using a synchronous three-pulse mode and asynchronous five-pulse mode, switching between these pulse modes can beperformed satisfactorily at an arbitrary boundary point shown by abroken line in FIG. 9.

There are the following advantages when control is performed bycombining pulse modes of different numbers of synchronous pulses andalso when there are a plurality of switching timings in one cycle of avoltage command.

For example, a pulse mode operation having equivalently changed thenumber of synchronous pulses can be achieved by continuously using pulsemodes of a different number of synchronous pulses at an appropriaterate. More specifically, when a synchronous three-pulse mode and asynchronous five-pulse mode are used at a rate of 1:1, for example,equivalently a four-pulse mode can be achieved from a viewpoint of thenumber of times of switching per unit time. In this case, by alternatelyusing a synchronous three-pulse mode and a synchronous five-pulse modefor each of the above sections, the reproduction precision can beimproved more than that when the pulse modes are switched at each cycleof a voltage command phase.

The rate of using synchronous pulse modes does not have to be the rateof 1:1 mentioned above, and any arbitrary rate can be used. When asynchronous three-pulse mode and a synchronous five-pulse mode are used,there are six sections as selectable sections in one cycle of a voltagecommand phase (see FIG. 9). With regard to the number of times of usinga synchronous three-pulse mode, seven ratios from zero to six times canbe selected. Furthermore, with regard to the selection pattern of bothpulse modes, a fixed pattern can be used such as a synchronousthree-pulse mode is selected two times at the beginning, and thenselecting a synchronous five-pulse mode one time, and repeating thispattern, or the pattern can be randomly selected by keeping a set usingrate.

In the carrier-wave comparison system or the phase reference systemexplained in the BACKGROUND ART section, a synchronous pulse mode isswitched in principle at each cycle of a voltage command phase. In thesesystems, it is possible to switch a synchronous pulse mode at pluraltimes in one cycle of a voltage command phase. However, this has a riskof lowering the reproduction precision considerably, generating a largechange in an output voltage at the time of switching a synchronous pulsemode, or causing unnecessary switching. Therefore, this switching methodcannot be a preferable control method.

Other advantages are as follows. That is, when a pulse mode istemporarily switched to a pulse mode of a large number of synchronouspulses to increase the voltage precision of an inverter, for example, itis not necessary to wait for a lapse of one cycle of a voltage commandphase, and this suppresses waste of time.

When a synchronous pulse mode is changed in a relatively long timerange, smooth switching can be performed at a high speed by graduallychanging a using rate of both pulse modes before and after switching.

As explained above, according to the controller of a power converter ofthe present embodiment, the controller can have a plurality ofswitchable timings in one cycle of a voltage command at the time ofusing a combination of pulse modes of different numbers of synchronouspulses. Therefore, other pulse modes can be performed in high precisionby combining plural synchronous pulse modes, and waste of time ofswitching synchronous pulse mode itself can be also suppressed.

In the first to fourth embodiments described above, switching patterncalculations have been explained for a case of controlling a two-levelthree-phase inverter in a synchronous three-pulse mode and a case ofcontrolling this inverter in a synchronous five-pulse mode. However, thepresent invention can be also applied to a multi-level inverter such asa three-level inverter, a multi-phase inverter other than three phases,and an inverter having a larger number of synchronous pulses. That is,according to the controller of a power converter of the aboveembodiments, the controller can be applied to any kind of a powerconverter that supplies an alternating-current voltage to a load byusing a synchronous PWM control.

INDUSTRIAL APPLICABILITY

As described above, the controller of a power converter according to thepresent invention is useful for suppressing an error between a voltagecommand and an inverter output voltage and for responding to a voltagecommand at a high speed.

1-6. (canceled)
 7. A controller of a power converter, applied to a powerconverter comprising an inverter that includes a plurality ofsemiconductor switching elements and controls the semiconductorswitching elements of the inverter by using a pulse width modulation,wherein the controller comprises: a voltage-command signal generatorthat generates a voltage command signal; and a switching patterncalculator that calculates a switching pattern to control thesemiconductor switching elements of the inverter based on the voltagecommand signal, and the switching pattern calculator calculates aswitching pattern of a synchronous PWM system, and outputs a switchingpattern in which an average value (an output-voltage average value) ofoutput voltages output from the inverter matches the voltage commandsignal.
 8. The controller of a power converter according to claim 7,wherein a value on a biaxial orthogonal-rotational-coordinate system isused for the voltage command signal and the output-voltage averagevalue.
 9. The controller of a power converter according to claim 8,wherein an average value of output-voltage in sections obtained bydividing the voltage command phase on a stationary coordinate systeminto x (x is a natural number) is used for the output-voltage averagevalue.
 10. The controller of a power converter according to claim 8,wherein a component in a voltage-command-signal vector direction on thebiaxial orthogonal-rotational-coordinate is used for the output-voltageaverage value.
 11. The controller of a power converter according toclaim 9, wherein a component in a voltage-command-signal vectordirection on the biaxial orthogonal-rotational-coordinate is used forthe output-voltage average value.
 12. The controller of a powerconverter according to claim 7, wherein when a switching pattern of thesynchronous PWM system is calculated, the switching pattern calculatorselects at least one synchronous pulse number from among a plurality ofsynchronous pulse numbers and performs the calculation while switchingthe selected synchronous pulse numbers.
 13. The controller of a powerconverter according to claim 8, wherein when a switching pattern of thesynchronous PWM system is calculated, the switching pattern calculatorselects at least one synchronous pulse number from among a plurality ofsynchronous pulse numbers and performs the calculation while switchingthe selected synchronous pulse numbers.
 14. The controller of a powerconverter according to claim 9, wherein when a switching pattern of thesynchronous PWM system is calculated, the switching pattern calculatorselects at least one synchronous pulse number from among a plurality ofsynchronous pulse numbers and performs the calculation while switchingthe selected synchronous pulse numbers.
 15. The controller of a powerconverter according to claim 10, wherein when a switching pattern of thesynchronous PWM system is calculated, the switching pattern calculatorselects at least one synchronous pulse number from among a plurality ofsynchronous pulse numbers and performs the calculation while switchingthe selected synchronous pulse numbers.
 16. The controller of a powerconverter according to claim 11, wherein when a switching pattern of thesynchronous PWM system is calculated, the switching pattern calculatorselects at least one synchronous pulse number from among a plurality ofsynchronous pulse numbers and performs the calculation while switchingthe selected synchronous pulse numbers.
 17. The controller of a powerconverter according to claim 12, wherein the switching patterncalculator has at least one timing of switching synchronous pulsenumbers in a phase section of the voltage command signal on a stationarycoordinate system.
 18. The controller of a power converter according toclaim 13, wherein the switching pattern calculator has at least onetiming of switching synchronous pulse numbers in a phase section of thevoltage command signal on a stationary coordinate system.
 19. Thecontroller of a power converter according to claim 14, wherein theswitching pattern calculator has at least one timing of switchingsynchronous pulse numbers in a phase section of the voltage commandsignal on a stationary coordinate system.
 20. The controller of a powerconverter according to claim 15, wherein the switching patterncalculator has at least one timing of switching synchronous pulsenumbers in a phase section of the voltage command signal on a stationarycoordinate system.
 21. The controller of a power converter according toclaim 16, wherein the switching pattern calculator has at least onetiming of switching synchronous pulse numbers in a phase section of thevoltage command signal on a stationary coordinate system.